IEEE Microwave Magazine - June 2015 - 36

Tuning a filter is even more
challenging than tuning a single
resonator structure such as an
antenna or a VCO.

FOM =-L (Df ) + 20 log (
FOMT =-L (Df ) + 20 log e

substrate used for the SIW cavity, respectively. The values of the effective length and width of the SIW cavity
can be found using
L eff = L -

d 2 + 0.1 d 2
0.95.b
L

(2)

d 2 + 0.1 d 2 ,
0.95.b
W

(3)

and
W eff = W -

respectively [30], where d is the wall vias' diameter
and b is the wall vias' spacing. Hence, the frequency
of operation in a cavity-backed SIW antenna depends
on both the slot and cavity dimensions. As a result,
conventional tuning methods [28], [32] applied to slot
antennas are not effective in designing SIW antennas as
they only change the slot's modes. Higher-order modes
are excited in a waveguide cavity and lead to a poor
upper-band rejection or the appearance of undesired
resonance frequencies in the tuned states. As a result,
these modes also need to be considered when designing
a tunable SIW antenna.
The resonance frequency of a cavity resonator can
be changed by changing its effective length, width, f r,
and n r, as shown by (1)-(3). However, changing these
parameters might not be practical. While changing the
permeability is used in one method, the other methods
use field perturbation and reactive loading inside the
cavity to tune the SIW antennas.

SIW Tunable VCOs
VCOs are used as the local oscillator source in most
radar and communication applications [33]. Some of the
driving parameters for VCOs are low phase noise, high
output power, low dc power, high tuning range, and
high harmonic suppression. It is well known that the
performance of a VCO is highly affected by the Q of the
passive resonator.
The high Q, power handling, and isolation of the
SIW cavity resonators in addition to their integrability capabilities sound very promising when designing
oscillators. However, due to the difficulties in tuning
these structures, they have not been used in VCO structures that often. As a result, a tuning method by which
SIW resonators can be tuned over a wide tuning range
while their quality factor variations are negligible is
very demanding.
To have a fair comparison between different SIW
VCOs, two figures of merit (FOM) and FOM with tuning range (FOMT) are used in this article as

f0
) - 10 log (P)
Df

(4)

f0
# FTR o - 10 log (P), (5)
10
Df

where L (Df ) is the phase noise at offset Df, f0 is the
oscillation frequency, P is the consumed dc power
used by the VCO, and FTR is the frequency tuning ratio
percentage. The standard FOM for VCOs is shown in
(4), and (5) factors in the tuning range as well.

SIW Tunable Filters
The significant demand for dynamic spectrum management in cognitive and software-defined radios
increases the need for RF/microwave devices that
can operate in different bands [34]-[37]. This means
that microwave filters, as one of the major blocks in
designing such systems, need to cover more bands.
Various solutions for such filters can be employed,
such as filter banks, multiband filters, and tunable/
reconfigurable filters. Although filter banks deliver
a very high-quality performance, a very large area
is needed to accommodate them in a system. Multiband filters suffer from isolation and crosstalk
problems and, thus, are less appealing compared
with tunable/reconfigurable filters. The ease of fabrication, high-power handling, high linearity, and
integrability of SIW filters with other sections of the
microwave system make them a good candidate for
high-performance microwave circuits. However, due
to their resonant nature, SIW filters are narrrowband
and highly selective.
Finding a way to tune SIW-based filters over a wide
tuning range while maintaining their high Q results
in high-performance, wide-range operating filters.
However, this has not been the case for most of the
proposed tuning methods so far. Either the tuning
range is very low or the Q of the resonator drops when
a wide tuning range is achieved, making it more challenging when designing such filters.
On the other hand, the BW and in-band loss of a
multipole filter are related to the coupling coefficient
and external Q factors. For example, for a two-pole filter, the required K 12 value can be extracted from fullwave simulations using [38]
K 12 =

f 12 - f 22
,
f 12 + f 22

(6)

where f1 and f2 are the two resonant peaks seen in
S 21 simulation results of the cavities coupled to each
other, also known as odd- and even-mode resonance
frequencies, respectively.
The external quality factor (Q e ) can be also extracted
using a singly loaded SIW cavity resonator. The Q e can
then be computed from 3-D full-wave simulations and
by employing the expression [38]

June 2015

Table of Contents for the Digital Edition of IEEE Microwave Magazine - June 2015